Communication network with control plane network

ABSTRACT

A communication network includes nodes communicating over a current generation network, such as 5G and some, or all of the nodes are also capable of communicating over a previous generation network, such as 4G for instance. A third node receives a first signal from a first node over the current generation network, a second signal that is a retransmission of the first signal from a second node over the current generation network, and network dimensional parameters from the first node and/or the second node over the previous generation network. The network dimensional parameters enable the third node to determine precise locations of the first node and the second node. Using a function of the network dimensional parameters, the third node correlates the first signal and the second signal to generate a simplified signal therefrom. A secondary wireless mesh acts as an overlay control plane via a non-shared non-virtualized external controller.

CROSS-REFERENCE TO RELATED APPLICATIONS

This application for patent is a Continuation in Part (CIP) of andclaims priority to U.S. patent application Ser. No. 15/798,243 filedOct. 30, 2017 which has been allowed for issue and claims priority toU.S. Provisional Patent Application No. 62/414,786, filed Oct. 30, 2016,which is incorporated herein by reference.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a diagram illustrating an example of a topology ofcommunication network having co-linear nodes and rectilinear nodes.

FIG. 2 is a block diagram illustrating an example of a node of a digitalInfinite Impulse Response (IIR) repeater network according to someembodiments.

FIG. 3 is a block diagram illustrating an example of a conventional nodeusing free space optics to transmit and receive signals.

FIG. 4 is a block diagram of nodes in a repetitive pattern.

FIG. 5 is a schematic block diagram depicting an example of a node withIIR filter implementation.

FIG. 6 is a schematic graph indicating an example of intensity of anoriginal signal propagating through multiple nodes in a digital-IIRrepeater network and intensity of copies of the original signaltransmitted from successive nodes receiving the original signal.

FIG. 7 is a schematic block diagram of an example of a PLL moduleaccording to some embodiments.

FIG. 8 is a schematic block diagram illustrating an example of modulesin a node that can be employed for out-of-band (OOB) controlplane/network.

FIG. 9 is a diagram illustrating an example of an MIMO architecture thatcan be implemented over a geographically dispersed area with adigital-IIR repeater network.

FIG. 10 is a diagram illustrating an example of an out-of-band (OOB) toMU-MIMO connection.

FIG. 11 is a diagram of an example of a MIMO mobile antenna with four2×2 modules.

FIG. 12 is a flowchart of an example of a method of using a controlplane network to provide network dimensional parameters to communicationnetwork nodes.

DETAILED DESCRIPTION

FIG. 1 is a diagram illustrating an example of a topology ofcommunication network 100 having co-linear nodes and rectilinear nodes.The communication network 100 comprises clusters 102 and 104 withthirteen nodes in each cluster and internet gateway node 106 to provideinternet access to the nodes. Typical deployment can have as many as twothousand or more nodes within each cluster, interconnected via wirelesslinks, where each node can be spaced apart a predefined distance, e.g.,fifty meters to two hundred and fifty meters.

In some embodiments, each of nodes A1-A13 and B1-B13 can be connected tostreet lamps, thereby fixing each node to a specific geographic locationwith a substantially regular interval. Because many modern streets arepositioned along straight lines and a substantially regular interval,the poles on the same street are generally positioned along a linearsignal path and a substantially regular interval. To leverage this idealplacement, nodes can be coupled to street poles along a linear signalpath of the regular interval. Thereby, nodes can be positionedcollinearly and transmit signals generated either by a directional RFantenna or other directional EM transmitter to each other. Thus, aself-backhauling capability is established that can be used for thenetwork of luminaries, for ancillary functions of a smart grid network,for metro mobile internet access, as well as backhauling capacity forcellular 4G networks, emerging 5G networks, and for other applications.There can also be other street lamps and other objects between any twonodes since poles are typically as close as twenty meters apart,depending on the lighting requirement of the city landscape.

As is illustrated, some nodes can be positioned along a linear signalpath. See, e.g., nodes B1-B6 of the cluster 104. In a specificimplementation, a millimeter-wave signal (e.g., ˜60 MHz) is repeatedfrom node to node. A node, such as node B6, may receive the transmittedmillimeter-wave signal from multiple nodes, such as node B5, B4, B3, B2,and/or B1. Assuming node B6 receives node B4's transmission, thereceived signal at node B5 and the received signal at node B6 mayslightly differ due to attenuation and the relative distances of nodesB5 and B6 from node B4. Node B6 will also receive a transmission of thesignal from node B5 as well. Since node B6 has two transmissions of thesignal (one transmission from node B4 and another transmission from nodeB5), node B6 can apply a correlation method to the received signals toobtain a strengthened, correlated signal and to reduce the interferenceby the multiple transmissions of the signal. If there are any othernodes interconnected to node B6 that have not received the signal, thennode B6 transmits the correlated signal to those other interconnectednodes. Thus, correlation is performed on the signal as a function of therespective distances of the interconnected nodes, signal processingdelay at each of the interconnected nodes, and the respectiveattenuation of the multiple transmissions. If those other interconnectednodes receive any of the other transmissions from node B4 and/or nodeB5, then those other interconnected nodes can also apply a correlationmethod to the multiple transmissions of the signal that is received.

Once node B5 receives a second signal, a symbol-time signal having oneor more bits per symbol that is transmitted from node B4, the secondsignal can be down converted. Depending on how a first signal wastransmitted, the second signal can be down-converted to baseband from asignal or can be converted from a signal to an electrical basebandsignal. The second signal can have attenuation due to free space loss,atmospheric absorption, foliage or other factors, and has a propagationdelay of tp1. Symbol-time second signal is then filtered, signalprocessed, and amplified (as a third signal with a time delay of tp1+Δp,where Δp is the signal processing delay of a node). The third signal isthen up-converted to radio frequency (and/or converted from anelectrical signal to an optical signal), and transmitted to otherinterconnected nodes (e.g., node B6).

Since nodes are co-linear, a fourth signal received at node B6 is acomposite of the first signal from node B4 (with additional signalattenuation and propagation delay of tp1+tp2) and the third signal fromnode B5 (with signal attenuation and propagation and processing delay oftp1+tp2+Δp). The fourth signal is processed with signal correlation atnode B6 to obtain a correlated fifth signal using signal processing(“SF”) techniques, similar to echo cancellation or other SP techniques.Ideally, correlation operates on an analog signal without conversion toa digital format since conversion to a digital format would increase thelatency delay.

Clusters 102 and 104 can have a total bandwidth of 1 Gbps or moreEthernet traffic. The topology of the communication network 100 is suchthat traffic originating from internet gateway node 106 travels to eachof the nodes of clusters 102 and 104. Each node of clusters 102 and 104repeats the analog signal originated by internet gateway node 106 toother interconnected nodes. Since the nodes do not store and forward anypackets, the latency to deliver data is extremely small. The activelinks in clusters 102 and 104 can be determined by a predefined method,for instance a spanning tree method or an alternative method, such thatthe mesh topology forms a tree structure without loops. Cluster 102 and104 can have a leaf-trunk convention, where a trunk direction is towardsinternet gateway node 106 and a leaf direction is towards a node.

A major criterion of the mesh network is that each node of clusters 102and 104 is able to receive data traffic from internet gateway node 106.One or more of the nodes A1-A13 and B1-13 should also be co-linear sothat some co-linear nodes can receive multiple transmissions of thatrepeated signal for correlation. Furthermore, nodes A1-A13 and B1-B13can be wirelessly or with-wire connected to end user devices via Wi-Fior via Ethernet with a single or multiple physical wire connections,such that the end users can access the internet via the internet gatewaynode 106.

A point-to-multipoint directional beam can also be used to reinforce thesignal that is transmitted from multiple co-linear and/or rectilinearnodes by repeating the same signal in a coherent manner. Each subsequenttransmission of the signal from node to node can be correlated andstrengthened to improve the overall range of the network and to reduceinterference by the multiple transmissions of the signal.

The mesh network can thereby be scaled with as many or more than 2,000nodes (which is a least 20 to 50 times the nodal size of conventionalWi-Fi mesh access point solution without the reduction of effectivetraffic payload speed associated with packet latency). This largenetwork size scaling can be attributed to quick repeating of anup-converted RF signal and/or free space optical signal to basebandelectrical signals and back to up-converted RF signal and/or free spaceoptical signal without the use of packet store and forwarding.

FIG. 2 is a block diagram illustrating an example of a node 200 of adigital Infinite Impulse Response (IIR) repeater network according tosome embodiments. The node 200 comprises transmitters Tx1-Tx4, receiversRx1-Rx4, and a matrix switch 202. The matrix switch 202 directs the dataflow for the transmitters Tx1-Tx4, the receivers Rx1-Rx4, and anyinternal node traffic. The internal node traffic can comprise telemetryinformation (including diagnostic data of the respective node and thenetwork) and external backhauling traffic from a Wi-Fi mesh access point(“AP”) or from a gigabit Ethernet port 1000 Base-T.

Since the mesh topology forms a tree structure without loops, the matrixswitch 202 can be reconfigured to orient the transmitters Tx1-Tx4 andthe receivers Rx1-Rx4 in accordance with the network topology and linkdirections. In some embodiments, the matrix switch 202 can be a logiccircuit that connects one of the receivers of a node to one of thetransmitters of that node to retransmit a signal that was received toone or more active links with other interconnected nodes. For instance,if the receiver Rx4 receives a signal that is to be retransmitted viaactive links that are supported by the transmitters Tx1 and Tx2, thenthe matrix switch 202 will route the received signal from the receiverRx4 to the transmitters Tx1 and Tx2 to be transmitted to other nodes viathe active links. Assuming the transmitter Tx3 does not support anactive link, then the transmitter Tx3 will not transmit the receivedsignal. Furthermore, the matrix switch 202 may also correlate allinbound traffic in the co-linear direction to retransmit a correlatedsignal to the other nodes.

FIG. 3 is a block diagram 300 illustrating an example of a conventionalnode using free space optics to transmit and receive signals. A node cancomprise optical transmitters Tx1-Tx4, optical receivers Rx1-Rx4, and amatrix switch 302 having a correlator processor 330. Here, the receiversRx1-Rx4 comprise logic blocks for optical to electrical conversion andthe transmitters Tx1-Tx4 comprise logic blocks for electrical to opticalconversion to generate the free space optical signal.

For the receiver Rx1, an optical detector 320 is used to detect a lasersignal from either an adjacent node and/or from a co-linear node beyondthe adjacent node. Once the optical signal is converted to an electricalsignal, the converted electrical signal is amplified by an amplifier 322and then filtered by a filter 324. The filtered signal is then input tothe matrix switch 302. The matrix switch 302 transmits the filteredsignal to downstream or upstream nodes using any of its transmittersafter pulse-shape filtering by a filter 332, and then converting thefiltered signal to an optical signal via an LED laser diode driver 334.The transmitters Tx1-Tx4 and the receivers Rx1-Rx4 can have similarhardware implementations and operate on the same optical wavelength.

Internal node generated traffic such as telemetry data and/or ancillarytraffic from video camera, audio microphones, and other devicesconnected to the node can be digitized and packetized according to IPprotocols. Those packets can be fed to a gigabit Ethernet layer 2 switch326 via a Wi-Fi logic block 328 or any other connection means. The layer2 switch 326 has controls from the correlator processor 330 from withinthe matrix switch 302 to throttle the first-in-first-out (“FIFO”) withinthe layer 2 switch 326 so that traffic collisions can be avoided.

Packet signals coming out of the layer 2 switch 326 are sent to themodulator/encoder+demodulator/decoder 329, which conforms to standardgigabit Ethernet packets. The digital signals are then modulated andencoded for transmission to any of the transmitters Tx1-Tx4 of the node.The direction of the digital signals can be determined by theconfiguration of the matrix switch 302. The correlator processor 330 cantime align multiple received signals from either a co-linear directionor rectilinear direction to form a strengthened, correlated signal andto reduce interference by the multiple transmissions of the signal. Thealignment is aided by information of the geo-location of the nodeswithin the cluster. The geo-location data can be used to determineco-linearity of neighboring nodes and the relative distances betweennodes. This information is used to determine the phase shift and signallevel of nodes to form a correlated, strengthened pulse signal and toreduce interference from the multiple transmissions. The correlatorprocessor 330 can also determine signals that are non-correlated andthus make a decision as to whether to send the correlated signal forwardor to decide that there is a collision during this pulse signaltimeframe.

In addition to the optical signals, electromagnetic waves of amillimeter-wave frequency range is employed for data communication. At amillimeter-wave frequency range (e.g., ˜60 GHz), even a path lengthdifference of 4 micron can lead to a one-degree phase shift for signalpropagation within a silicon substrate (of which dielectric permeabilityis about 11.9) of a repeater node. This would not be a major issue if ananalogue repeater, including an on-chip antenna for reception and anon-chip antenna for transmission, were implemented within the samesubstrate (same chip) of the repeater node, because it is possible tocontrol a trace length between the on-chip antennas with high enoughprecision to ensure phase coherency for proper repeater operation.However, this would violate a requirement that an RX (received) signaland an amplified TX (transmitted) signal be isolated from each other byat least 100 dB to prevent an aversive positive feedback, which wouldsaturate the receiver's low-noise amplifier (LNA).

In a multi-chip implementation where the RX and TX sides of the repeaterare embedded in distinct chips with a physical separation that is largeenough to provide enough RF shielding to satisfy the 100 dB isolationrequirement would lead to uncontrolled phase uncertainty as well as afrequency-dependent propagation delay up to 1 ns. In a typicalsubcarrier (OFDM) modulation scheme, the maximum propagation delay in along repeater chain (5 or more repeater nodes) could be comparable tothe symbol length for multi-gigabit data transmission, which results ina severe distortion/degradation of the combined signals, requiring useof sophisticated time domain equalization to recover the original signalon a receiving end. In addition, the delay and delay spread are furtherimpacted by inevitable reflections by various impedance mismatches anddiscontinuities, especially where a feed line is connected to theon-chip antenna. Even for a good antenna Voltage Standing Wave Ratio(VSWR) of 1.5, about 20% of the signal voltage, or 4% of the signalpower is reflected. Although a 4% reflected power at the feed point doesnot seriously reduce the antenna transmission efficiency, the reflectedwave will couple to the power amplifier (PA) of the transmitter todistort the output (of the PA). The resulting reverberation will greatlyincrease the unwanted delay spread.

In a dense repeater deployment, an inter-node spacing could be as smallas 40 m, and each copy (phase warped and time delayed) of an originalsignal will have a similar magnitude. This makes use of a Finite ImpulseResponse (FIR) equalization filter ineffective, which in turn mandatesemployment of an adaptive IIR tap-delayed equalization filter. However,such a filter cannot be easily implemented in a strictly analogue waybecause while the IIR equalization filter itself can be done entirely inan analogue domain, the adaptive algorithm of the IIR equalizationfilter must be done in a digital domain, which would entail use ofultra-high-speed analogue-to-digital converter (ADC). The analogue IIRequalizer is also not suited for tap length of more than 10 nodesbecause of its intrinsic numerical instability. The complexity of theanalogue IIR filter is further compounded by the fact that at least twotaps for each network node in a chain are needed to accurately model apropagation delay spreads created by each node.

In view of the issues discussed above, various embodiments describedherein are directed to provide configuration of a digital-IIR repeaternetwork applicable for communicating signals, such as millimeter-wavesignals, between nodes of the digital-IIR repeater network.Millimeter-wave signals have strong directivity and applicable toshort-range communication, such as the communication between streetlumps. Millimeter-wave signals in this paper are intended to representsignals in a frequency range of 20-100 GHz, for example about 60 GHz,which is a license free frequency range. For fundamental architecture ofthe digital-IIR repeater network, applicable architecture is disclosedin U.S. Pat. No. 9,094,119, the entire contents of which areincorporated herein by reference.

1. Digital Infinite Impulse Response (IIR) Approach

According to some embodiments, to implement a digital equalizationscheme, an IIR filter is implemented in the digital domain, after A-to-D(analog to digital) conversion. Further, to achieve the digitalequalization scheme, a direct conversion (zero IF) architecture may beemployed, because this architecture can be a most feasible frequencyconversion architecture suited for a millimeter-wave frequency range. Inthe direct conversion architecture, output of an ADC can be already at abaseband (BB) level, and therefore it is feasible to use a digitalsignal processor (DSP) to implement a digital IIR filter. Thedigital-IIR coefficients can be specified by their respective amplitudesand phases. If we assume that the minimum distance between two meshnodes is 20 m, then if the geographical location of the mesh networknodes can be determined to within 5 cm, then the amplitude of the IIRcoefficients can be estimated to within a +/−0.5% accuracy, which wouldbe more than sufficient even for a large mesh network. For a shortnetwork, the phase of the mesh node IIR coefficients can't be reliablydetermined by geolocations alone, but accuracy can be increased inreal-time using the phase and symbol clock synchronization mechanismdescribed later in this paper.

FIG. 4 is a block diagram 400 of nodes in a repetitive placementpattern. In the diagram 400, nodes 402-1 to 402-9 (collectively, thenodes 402) are intended to represent a repetitive placement pattern. Therepetitive pattern in the example of FIG. 4 is square, but otherrepetitive patterns, such as rectangular, hexagonal, or the like, can beused. A node generating a simplified signal must be aware of the form ofthe repetitive placement pattern to utilize certain simplificationtechniques, such as IIR filtering.

In the example of FIG. 4 , the node 404 is intended to represent a nodethat correlates a plurality of signals from a corresponding plurality ofnodes (a subplurality of the nodes 402) when the plurality of signalsare all a transmission or re-transmission of a signal from a singlesource node (e.g., a wireless device coupled to the mesh via a wirelesslink, a gateway node, or the like). The node 404 generates a simplifiedsignal generated using a function of respective distances of thesubplurality of the nodes 402 in the repetitive placement pattern. Thefunction may also take into account signal processing delay at thesubplurality of nodes 402 and attenuation associated with the signaltransmission from the subplurality of nodes 402.

The arrow 406 is intended to represent a simplified signal correspondingto correlation of the plurality of signals from the subplurality of thenodes 402. It may be noted, the lines connecting the nodes 402 with oneanother and with the node 404 indicate adjacency, but it should beunderstood the nodes 402 and the node 404 may be within range of othernodes that are farther away.

FIG. 5 is a schematic block diagram 500 depicting an example of a nodewith IIR filter implementation. The node includes an Rx antenna 502, anRx co-linear section 504, a DSP 506, a Tx co-linear section 508, and aTx antenna 510. The Rx co-linear section 504 includes analog-digitalconverters 514 and a voltage controlled oscillator 516, and the Txco-linear section 508 includes a voltage controlled oscillator 518. Amillimeter-wave signal received by the RX antenna 502 isanalog-to-digital converted by the Rx co-linear section 504, and aconverted digital signal is processed by the DSP 506, where an IIRfilter 512 of the DSP 506 performs co-phasing of an output signal to theinput signal, and the output signal is digital-to-analog converted bythe Tx co-linear section 506, and output from the Tx antenna 508.

Advantageously, since the DSP 506 is already employed for a basebandmodem, the digital implementation of the IIR filter 512 comes at almostno cost. By contrast, an analogue implementation of an IIR filter may benecessarily bulky and expensive, and may need an additional set of ahigh-speed ADC's to compute the adaptive coefficients for the IIRfilter, which greatly adds to the cost of the analogue implementation ofthe IIR filter.

Furthermore, because of processing delay of computation by the DSP 506,each copy of an original millimeter-wave signal (e.g., input signal) issufficiently separated from each other and from the originalmillimeter-wave signal. For that reason, the copies are less likely tointerfere with each other, and the delay spread of each copy is lesslikely to accumulate from one node to the next node. If, furthermore,nodes within a linear chain are precisely and equally spaced, whichimplies that the copies (and the original signal) are also precisely andequally spaced in time, then a simple tapped delay line architecture forthe IIR filter 512 with the delay that is equal to the time spacingbetween two consecutive nodes would remove most of temporal signalcorrelations. This assumes that the received signal is not directlyretransmitted since that would imply retransmission of all copies of theoriginal millimeter-wave signal emanated by all nodes that came beforethe current node which would greatly complicate the equalization processbecause the continuous accumulation of the successive copies would causethe IIR filter 512 to become more and more unstable. It would also causethe delay spread to accumulate, thereby forcing the IIR filter 512 todeal with the ever increasing delay spread, thus requiring a much morecomplex IIR architecture to deal with both the processing delays and thedelay spreads.

Moreover, by retransmitting the IIR filtered signal rather than thereceived signal (input signal), the combined waveform takes on a cleandecaying comb shape, which is much more amenable to the simple tap delayline architecture discussed above, at least in theory. However, thecombination of delay spread, delay jitter, and high environmental noise(thermal plus flicker noises) and interferences would corrupt thetransmitted signal if the total path-length is large. FIG. 6 is aschematic graph 600 indicating an example of intensity of an originalsignal propagating through multiple nodes in a digital-IIR repeaternetwork and intensity of copies of the original signal transmitted fromsuccessive nodes receiving the original signal.

To reduce cumulative signal corruption, each IIR filtered signalgenerated at node 1-5 is further regenerated on its symbol level afterdemodulation and then reshaped and re-modulated for retransmission. Aslong as the symbol level decision (slicing) is correct, which requires alow bit error rate (BER), the regenerated signal will be close to thetransmitted original millimeter-wave signal from node 0, and theretransmitted copies will be largely unaffected by signal corruption.Since the digital-domain IIR filtering, and subsequent demodulation andsymbol quantization (or more generally, symbol decision where thedecision process could be hard such as vector quantization, or soft likemaximum likelihood decision whose output is forwarded to a soft forwarderror correction (FEC)) are already a part of the baseband decodingprocess, the regenerated symbol can be tapped directly from the outputstream of the relevant stage of the baseband demodulation and decodingprocess. In effect, the symbol regeneration is free for the taking, andadditional DSP processing may not be needed. The advantage of using theregenerated symbol for retransmission is that the regeneration processlargely removes signal distortion and corruption due to noise,interference, delay spread, etc. as long as the symbols are correctlydetected, which is to a large extent true in a very low BER environment.The regeneration improves BER for all downstream nodes by preventingdownstream error propagation. However, a symbol decision error willpropagate downstream, which has the potential to increase BER for alldownstream nodes. Thus, like all decision-based equalization techniques,the digital-domain IIR equalization discussed in this paper is anonlinear equalization technique and exhibits extremely a nonlinearbehavior. On the other hand, even though a well-constructed linearequalizer can reduce inter-symbol interference (ISI), the regenerativeequalization may actually cause worse error propagation when the BER issufficiently high.

At a millimeter-wave frequency range (e.g., ˜60 GHz), even a relativelysmall phased array can have an extremely narrow beam width. This leadsto two important simplifications. First, only a single relay chain needsto be considered for the IIR process since the interference fromadjacent linear chain is practically negligible. Second, the very narrowbeam width and the negligible contribution from diffraction effect meanthat multi-path fading can be ignored. This makes determination ofmagnitudes of IIR coefficients extremely simple because the magnitudesfollow from the inverse square law of the free space propagationmultiplied by the exponential decay characteristics of atmosphericabsorption. The atmospheric absorption has a small dependence on thewater vapor content in the atmosphere and needs to be monitored toobtain the IIR coefficients in real-time. The phases of the IIRcoefficients, however, are extremely sensitive function of the exactdistance between the transmitting and receiving nodes, since even a 2.5mm uncertainty in the distance between network nodes could mean a totalphase uncertainty even though any possible uncertainty in the magnitudeof the IIR coefficient is entirely negligible.

The complex angles associated with the tap weights of the IIR filterbecome identically zero when the retransmitted signal is exactly inphase with the phase of the free propagating carrier wave. This can beaccomplished with a carrier phase acquisition phase to lock the localoscillator's phase (and frequency), and then using the phase-lockedlocal oscillator to modulate the signal to be retransmitted. However,care must be given to account for the phase difference between themodulation stage and the transmitting antenna, as well as the differencebetween the free propagating carrier wave at the receiving andtransmitting antenna. This would be less of a problem if the RX and TXantennas and RF electronics are all embedded within the same substrate.However, the 100 dB isolation requirement between RX and TX may renderit unrealizable.

It is clear that carrier phase acquisition, or recovery, is essentiallyfor success of the aforementioned method. Carrier recovery together withsymbol-clock recovery must be correct in order to demodulate the symbolsand recover the transmitted information. A symbol clock could be at theright frequency, but at the wrong phase the demodulation would still beunsuccessful. However, millimeter-wave devices tend to have much higherphase noises which makes phase and clock recoveries difficult.Fortunately, due to recent advances in millimeter wave technologies,existing millimeter-wave frequency range (e.g., ˜60 GHz) transceiversare already equipped with fairly accurate carrier recovery mechanism toenable the receiver to properly demodulate higher quadrature amplitudemodulation (QAM) signals. QAM demodulation requires accurate frequencyand phase synchronizations to properly align I and Q axes. Any error incarrier frequency estimation would cause the received signalconstellation to rotate continuously in the receiver's 1-Q plane, andthe estimation error would rotate the received signal constellationrelative to the assumed I-Q axes. Higher QAM such as 32 QAM or 64 QAMhave relatively large constellations, hence even a small phase errorwould lead to a large error vector, which makes technologies attained bysuch transceivers particularly remarkable.

In view of the above issues, in some embodiments, to ensure that theretransmitted signal and the received signal are in phase with eachother, the RX chip and the TX chip are placed on the same plane andintegrating the respective antenna with the RX and TX chips.

In some embodiments, to ensure the in-phase state of the retransmittedsignal and the received signal, the same local oscillator (LO) is usedfor modulation and demodulation. When a second LO (e.g., the VCO 518) isemployed for TX, while keeping the RX LO (e.g., the VCO 516) the same,then it would be possible to use the TX LO to synchronize the phasesbetween RX and TX. To ensure proper phase synchronization, the phasedetector for the second LO measures the relative phase between the RXsignal and the TX signal measured at their respective center feedpoints, and the phase error signal (from the second phase detector) isused to control the VCO through the normal phase locked loop (PLL)mechanism. The main difference between the above approach and atraditional PLL approach is, instead of monitoring the relative phasebetween the external clock signal and the LO clock signal, monitoringthe relative phase between the input clock signal and the output signalwhere the output signal is driven by the second LO. It is noted that thephase shift between the 2nd LO clock signal and the output signal isalready accounted for in the phase error signal, and therefore there isno need to estimate the phase compensation needed for the desired signalsynchronization.

In some embodiments, to ensure the in-phase state of the retransmittedsignal and the received signal, a compensation phase of each node isiteratively determined by performing a linear search (1-D search) firstfor the second node (node 1) which receives a beacon signal from thefirst node (node 0), and then the compensation phase of the third node(node 2) after the phase of the second node has been determined, and soon. Advantageously, linear search is a very efficient well establishedtechnique, which can find a fairly accurate answer typically within afew tries. The objective function used with each iteration can be takento be the signal correlation after decimation to reduce computationrequirement.

Once the in-phase requirement is satisfied for every node in apropagation chain, all nodes can retransmit coherently, which in effectturns those nodes into a giant one-dimensional phased array. The IIRweights now become: (1), (1, ½), (1, ½, ⅓), (1, ½, ⅓, . . . , 1/n), . .. if atmospheric absorption is ignored, or, more generally: (1), (1,a/2), (1, a/2, a²/3), . . . (1, . . . , a^(n−1)/n) since the IIRcoefficients are known except for an atmospheric absorption factor,which can be easily measured, any additional adaptation, or learning, isunnecessary so long as all nodes are within light of sight (LOS). If anyparticular node or nodes are offline, the corresponding IIR weight(s)can simply be set equal to zero. It can be shown that all such IIRfilters are numerically stable, irrespective of how many nodes are inthe linear chain. This avoids a complicated process of root manipulationto ensure stability of the adaptive IIR equalization filter which makesit difficult to extend the adaptive approach to much more than a dozenof nodes.

The simplicity of the IIR equalization approach is also amenable to fastDSP computation which can drastically reduce computational load of aDSP. For cases where LOS conditions cannot be completely satisfied, anadaptive FIR (finite impulse response) filter could be used inconjunction with the aforementioned IIR filter to perform partiallyadaptive multi-node equalization. Such adaptive approach does not incurstability issues since FIR filter is always stable.

It should be noted here that if the nodes are not substantiallyregularly spaced, then the corresponding digital-domain IIR filter maybecome far more complex, requiring far more tap weights than the numberof nodes. For the same reason, if the phase synchronization is notenforced, then the corresponding digital-domain IIR filters may becomefar more complicated, requiring complex adaptive approach to determinethe complex phases of the tap weights. This is especially true when theposition of each node cannot be maintained to within a millimeter sinceeven a 2 mm movement can lead to a phase shift of 144°.

In a way, the digital-domain IIR equalization process is similar to howa RAKE receiver works in code-division multiple access (CDMA). In CDMA,a RAKE receiver attempts to collect dominant time-shifted copies of theoriginal signal by providing a separate correlation receiver for each ofthe multipath signals. The correlator outputs are then time-shifted andcombined to achieve enhanced signal by coherently aggregate and mergethe signal energy associated with each time-shifted version to improvethe signal to noise ratio (SNR). The gain in SNR is called a RAKE gain.Although the propagation environment considered for this invention hasnothing to do with multi-path fading, the coherent radiation from eachnode can still be accurately described as a time-shifted copy of theoriginal signal, and the phase locking mechanism for each node issimilar to the correlation mechanism, because the relative phaseinformation from a RAKE finger is contained within its output. Inaddition, although the coherent combining of the RAKE outputs is moreakin to a FIR filtering, any false image of the FIR mechanism resultingfrom the RAKE operation is largely removed by the resulting codecorrelation action. For a multi-gigabit 60 GHz system, however, anyparallel to a RAKE operation will not work well in practice for two mainreasons: First, any possible spreading ratio would be too low tosuppress ISI effectively. Second, with a peak symbol rate reaching 500MSPS (mega-symbol per second), the RAKE correlation search operationswould be simply too costly given the current status of DSP technology.By placing the network nodes as equally spaced lattice points, each nodewill see a superposition of regularly time-shifted versions of anoriginal signal from all nodes in its upstream. This greatly simplifiesthe equalization operation (one can logically consider the RAKEoperation to be an equalization operation in CDMA flavor) because of theregularity of the time-shifted superposition wherein even the amplituderatios are already known in advance.

The gain from the coherent combining of the time-shifted signals can beconsidered to be the analogue of a RAKE gain, but is more appropriatelylooked upon as an array gain, and the linear chain is nothing but aversion of synthetic aperture array (or more simply, aperture array, forshort). This array gain effectively increases the single antenna gain byan array factor, which, for far downstream nodes, could exceed 4 dB. Thehigher effective antenna gain also translates into narrower beam width,which further reduces any multi-path effects.

2. Clocking Distribution, Slips, and Synchronization

Ideally, carrier recovery and symbol clock synchronization should keepevery node within a linear node chain fully synchronized (carrierfrequency and symbol timing) and appropriately phased. However, since aPLL relies only on a loop filter for averaged phase detection errors,the PLL may not sufficiently prevent clock slips. FIG. 7 is a schematicblock diagram 700 of an example of a PLL module according to someembodiments. The PLL module in FIG. 7 includes a phase detector 702, aslicer 704, a loop filter 706, and a unit time delay 708. In operation,a signal input to the phase detector 702 is output therefrom and theoutput signal is combined with an output of the unit time delay 708 andthe combined signal is output to the slicer 704. Input to and outputfrom the slicer 704 are combined, and the combined signal is input tothe loop filter 706. Then, an output of the loop filter 706 is combinedwith the unit time delay 708, and the sum is provided to as to becombined with the output of the phase detector 702.

According to some embodiments, the PLL module 700 does not actuallydetect a frequency of a signal directly. Instead, the phase is directlydetected by the PLL module 700, and the frequency is inferred from thedetected phase. Furthermore, the relation between frequency and phasemust satisfy Heisenberg's uncertainty principle in that they cannot beboth defined to arbitrary accuracy. On the other hand, any minutedifference in frequencies between a LO and the reference clock sourcewill cause a phase error to increase over time until the negativefeedback voltage controlling the VCO begin to kick in to pull the LOfrequency closer to the reference frequency. The exact phase/frequencydynamics depends on both the feedback gain and the bandwidth of the loopfilter 706. A low feedback gain together with a low corner frequency forthe loop filter 706 may cause the inferred frequency to be close to thereference frequency, but at the same time also allows for higher clockslippage rate. On the other hand, a larger feedback gain and a widerloop bandwidth may provide less frequent slippages but also largerfrequency excursions called reference spurs because the loop filter 706no longer is able to attenuate the FM (frequency modulation) likeexcursions.

In order to maintain phase coherencies among multiple nodes, it would beunwise to allow those nodes to have its own LO clock and rely only oncarrier phase recovery for synchronization (symbol timing recovery isless troublesome since even a 100 pico-second clock jitter will havelittle impact on symbol timing). For example, if each of the multiplenode uses a crystal clock source frequency multiplied to serve as themaster clock, then the synchronization process would pit one clockagainst another. Even when the PLL module 700 is able to lock the twoclocks well enough, the static phase difference will be large unless thetwo crystals happen to behave exactly the same way. With a large staticphase error, the phase margin becomes inadequate to minimize slippagesand spurs. The situation becomes even more severe when multiple clocksneed to be synchronized.

In order to address this issue, in some embodiments, a satellite GPS(global positioning satelite) clock signal is used as a master clocksignal for the first node (node 0), and the remaining nodes (node 1-)primarily rely on carrier recovery to synchronize with the upstreamclocks. The downstream nodes may still keep track of the GPS clocksignal, but fall back on their GPS signals when a glitch occurs and thepreferred carrier synchronization fails or when there is a sufficientlylarge clocking discrepancy so that all nodes should be notifiedimmediately and coordinated for re-clocking. According to the use of theGPS clock signal selectively by the initial node, the downstream nodeswill not “fight” the master clock with their own local masters, andtherefore the clock slips as well as static phase errors can besignificantly reduced.

3. Out-of-Band Control Plane

An out-of-band network can be an earlier generation network, which the4G network is soon to be, with sufficient data transport capacity torelay information to mesh network nodes for real-time or time-varyingcoefficients and other network dimensional parameters enabling arelatively precise determination of the location of the mesh networknodes. Relatively precise is intended to mean having greater accuracythan GPS or within 10 cm.

The digital-IIR repeater network as described above is best suited forultra-high-speed data transmission efficiently. FIG. 8 is a schematicblock diagram 800 illustrating an example of modules in a node that canbe employed for out-of-band (00B) control plane/network.

During normal operations, the high-speed data is transmitted and relayedin one direction for a single data communication sequence. In order tocoordinate nodes in the digital-IIR repeater network to perform thephase and symbol clock synchronization described above, and otherroutine and emergency management tasks, the digital-IIR repeater networkmay dedicate a certain percentage of the data bandwidth to suchmanagement tasks either in a time division or frequency divisionfashion. In the alternative, the digital-IIR repeater network could relyon a much slower out-of-band overlaying network to perform thesemanagement tasks. An issue with the in-band control plane is that thenode requires the in-band channels to be able to reach other nodes inthe (two-dimensional) network, which means that more complex hardwareand firmware are needed to provide multi-way direct communications.However, because the millimeter-wave frequency range (e.g., ˜60 GHz) ofthe digital-IIR repeater network is limited, in order to communicate toa distant node, some sort of relaying is still needed, which furthercomplicates the issue.

Another issue with the in-band control plane approach is that eventhough the digital-IIR repeater network can be designed for faulttolerance by enabling it to reroute the relaying path if some linksbecome inoperative, such self-healing capacity is inherently limited andthus cannot cope with a severely impaired network where a fraction ofthe upstream nodes become inoperative at the same time. When thissimultaneous inoperability of upstream nodes happens, the in-bandcontrol also gets knocked offline, and thus means of communicating andcoordinating with the digital-IIR repeater network may be lost. Theinability to monitor, coordinate, and control the underlying digital-IIRrepeater network would essentially force communication offline.

In order to address these issues, in some embodiments, a lower-frequencylonger-range wireless network (secondary wireless network) is used as anoverlay network for the digital-IIR repeater network (primary wirelessnetwork). This out-of-band control network, optimally designed as aseparate mesh network with much higher degree of interconnectivity, canprovide a much higher resiliency and availability so that theout-of-band control network can maintain full connectivity even when thedigital-IIR repeater network is severely impaired.

Such an out-of-band control network (secondary wireless network) doesnot need to have high-speed capability if only monitoring and controlfunctions are needed. In a specific implementation, however, it would bedesirable to have a peak link speed in excess of 100 Mbps, so that theout-of-band control network can take over the data transport to providea temporally bridge for a small subset of links in the digital-IRrepeater network when the digital-IIR repeater network is so damagedthat it can no longer find a new route to completely restore datanetwork connectivity for the impaired old route. Bridging by thelower-speed out-of-band control network can drastically cut down on datanetwork throughputs along certain route(s). Even though a portion of thenetwork clients would see a huge drop in data speeds, it would still bebetter than if their data network become totally inaccessible.

4. Millimeter-Wave Frequency Range MIMO and Mobile Multiple AccessApplications

MIMO (multiple-input-multiple-output) architecture is typically designedfor lower frequency RF bands (e.g., UHF, SHF). According to MIMO, MIMOprovides enhanced communications performance on multiple manners,including as beam forming, which improves effective antenna gain andreduces mutual interferences, same frequency reuse, which createsmultiple parallel pipes using the same frequency to multiply thecapacity of a radio link, and interference cancelling, which creates anull or a low antenna cone along a particular direction to suppress thestrong interference coming from the particular direction. FIG. 9 is aschematic diagram 900 illustrating an example of an MIMO architecturethat can be implemented over a geographically dispersed area with therepeater network. The MIMO architecture in FIG. 9 includes a repeaternetwork including a node 902 a, which is connected to a gateway, aplurality of repeater nodes 902 b, and a wireless mobile unit 904carried by a user. The wireless mobile unit 904 has multiple antennasthat are configured to communicate with different nodes in the repeaternetwork.

More advanced MIMO such as MU-MIMO (multi-user MIMO) can furthermultiply the radio link capacity by using the so-called spatial divisionMIMO or spatial division multiple access (SDMA) which enables users totransmit signals at the same time and frequency to communicate to thesame base station at the same time by taking advantage of their uniquespatial signatures.

Most MIMO technologies are applicable primarily in high multi-pathfading environment. This is because in a MIMO system, amultiple-antenna-equipped transmitter (e.g., the wireless mobile unit904) sends multiple streams through its multiple built-in antennas, andon the receiving end (e.g., the repeater network) likewise is equippedwith multiple receiving antennas. If the number of transmitting antennasis N_(I) and the number of receiving antennas is N_(O), then the channelpropagation matrix, which characterizes the N_(I)×N_(O) propagationpaths, can be expressed as a N_(I)×N_(O) matrix. In order for the MIMOto work effectively, the conditionality of the channel matrix plays animportant role. The channel matrix is well-conditioned when the squarematrix formed by the product of N_(I) and N_(O) with its hermitianconjugate has at the least a number of eigenvalues which are finite (notexponentially small). For those eigenvalues, the correspondingeigenvectors can be considered space-time codes which can be used totransmit data; the same way that CDMA uses codes to transmit data. Themain difference, aside from the fact that the codes in CDMA are temporalcodes whereas in MIMO the codes are spatial-temporal codes, is thatthose eigen-codes in MIMO are theoretically mutually orthogonal and aretherefore are not subjected to inter-code interference which severelylimits CDMA's data capacity.

The efficiency of the MIMO in carrying data can be seen from thefollowing example. Here, it is assumed that a 4×4 MIMO is establishedwith 4 TX antennas and 4 RX antennas. If the channel matrix for this 4×4system is reasonably well conditioned so that 3 of its eigenvalues arefinite, then it would be possible to create 3 orthogonal channels (or“virtual wires”) using the same frequency band, for a 3 times frequencyreuse. In general, the magnitude of the eigenvalue is directly relatedto the free-space loss associated with its eigen-code, wherein if aparticular eigenvalue is so small that the final SNR is well belowunity, then it would not be useful for it to transmit data.

In a typical MIMO implementation, both the multiple TX antennas and themultiple RX antennas are closely spaced. This is especially true for amobile handset where there simply is not a lot of space to accommodatemultiple antennas except to pack them closely. If the antenna spacing ismuch smaller than the distance between the TX site and the RX site, thenin a LOS environment, all the channel matrix elements are almostidentical in values. Under these conditions, the channel matrix may beextremely ill-conditioned and there may be only one dominant eigenvalue,which means that both the TX antennas and the RX antennas may simply actlike a single antenna (with array gain). However, in an environmentwhere there are no dominant LOS path between the TX device and the RXdevice, then the individual propagation, even coming from two nearbyantennas, can take very different paths, hence the individual elementsof the corresponding channel matrix can be quite random. Such matrix istypically well-conditioned and can support multiple orthogonal channels.

For the millimeter frequency range (e.g., ˜60 GHz) of the repeaternetwork, both diffraction and reflection may play negligible roles indata propagation. The diffraction and reflection, together with thenarrow beam width, essentially rules out the contribution of multi-pathfading to the channel matrix. As such, it would be impossible to useMIMO the same way as it is used in lower frequency bands to enhance datacapacity. However, it would be feasible to take advantage of thesynthetic aperture array characteristics of the repeater network totransmit and receive signals from multiple nodes in the spatial array.In this scenario, it is no longer true that the antenna spacing is muchsmaller than the distance between TX and RX. In fact, in general, thereverse is true. Now imagine that the aperture array as a whole is theequivalence of a very large multiple antenna base station, then there isstill the question of whether a small mobile handset will have enoughspace to accommodate a plurality of tiny antennas so that they all havedistinct characteristics. If all handset antennas have almost identicalRF characteristics, then they are mathematically equivalent to a singlemobile antenna, in which case, it would correspond to a N×1 or 1×N MIMOarchitecture, which would permit beam forming, but no multiple spatiallyorthogonal channels are created, hence no SDMA.

According to some embodiments, in order to address this issue, antennasof the wireless mobile unit 904 that is configured to connect to therepeater network are multi-directed whereas nodes 902 a and 902 b of therepeater network are used as multiple antennas of a synthetic-wide areabase station in the MIMO architecture as depicted in FIG. 9 . In aspecific implementation, for the antennas of the wireless mobile unit904, instead of using a single 4×4 planar phased array, which wouldprovide a narrow beam width in a normal direction, a multitude of 2×2planar arrays are used with each planar array being directed to adifferent direction.

FIG. 11 is a diagram 1100 of an example of a MIMO mobile antenna withfour 2×2 modules. In the example of FIG. 11 , each 2×2 module includesfour patch antennas 1106 interconnected by micro-strip feeds 1104 andmicro-strip feed points 1102. By changing the feed points, each patchmodule will have its main lobe pointed at a different direction owing tothe differing phase relationships among the individual patches within asingle patch module. It is also possible to create circular polarizationfor each 2×2 module by employing two orthogonal feeds per module insteadof a single feed per module as shown in the example of FIG. 11 . For 60GHz bands, each patch measures around 3 mm×3.3 mm, which is roughly halfthe EM wavelength of the dielectric substrate. The feed lines are of theorder of a wavelength but the lengths can be adjusted so the requisitephase differences between patches are maintained.

In a specific implementation, the directional angles are separated bythe half-beam width of the individual 2×2 phased array, so that there isconsiderable overlap in solid angle between any adjacent phased arrays.Furthermore, the linear polarization vectors between adjacent phasedarray are also made to be nearly perpendicular to each other, and thenit becomes obvious that different phased array will tune to differentdirection of propagation as well as the direction of polarization,wherein the channel matrix is rendered well-conditioned to permitefficient SDMA.

It may not be sufficient to rely only on antennas with multipledirectional normal or polarization to ensure a well-condition behavior.For example, if one were to use only a single node equipped withmulti-directional antennas for the base station instead of a multitudeof nodes, then it can be shown that the resulting channel matrix can befactorized, which makes the matrix highly singular and ill-conditioned.Factorization is not possible when there are more than one node for thebase station.

An additional benefit of the mobile antenna configuration in theembodiments is that it can transmit/receive millimeter waves from a widerange of directions, in sharp contrast with a single 4×4 phased arraythat has higher gain but can receive signal in only a very narrowdirection. Although each 2×2 array has far less gain (and thereforewider beam angle), the combined MIMO gain more than compensate for thereduction in single antenna gain. It is apparent that a particularimplementation of the mobile antenna configuration in the embodimentscan overcome one of the biggest obstacles in applying themillimeter-wave frequency range for mobile applications. For lowerfrequency bands, where free space path losses are far smaller, thewireless mobile devices typically are equipped with low-gainomnidirectional antennas. The employment of multiple small phased arraysalso provides high antenna diversity and redundancy not found in currentcellular mobile handsets.

Another major issue to apply the millimeter-wave frequency range (e.g.,˜60 GHz) of the repeater network to mobile communications is the need toconstantly stay within LOS of the base station antennas to connect. Insome embodiments, this issue is solved by having a dense repeaternetwork where each node within the network comprises a fair number ofhigh gain phased arrays pointing every which way. This provides a densecoverage in such a way that any mobile handset will be at the LOS pathof at least a few nodes. In fact, since all those radio beams share thesame frequency, it is almost as if those multitude of beams convergingon (or diverging from) a particular handset came from a high multi-pathtype of diffraction/reflection. This makes ubiquitous coverage possiblefor the millimeter-wave frequency range.

5. Multiple Access in Millimeter-Wave Frequency Range by Beam Hoppingand Arrival Angle Estimation

Another major advantage of employing the repeater network as multipleantennas of a synthetic wide area base station is that the repeaternetwork already takes care of a ultra-speed backbone needed to feed datato individual nodes. A further advantage is that all nodes in thenetwork already have well-synchronized clocks, which makes it feasibleto produce coherent beam from disparate nodes. It should be remarkedhere that the repeater carrier phase synchronization primarily providesa reference phase for each node to facilitate high-speed data repeatingoperation. For beam forming and SDMA operations, an offset phase mayneed to be introduced for each antenna for each intended beam direction.The beam can be digitally steered simply by changing the offset phasefor each antenna (within a single node). To enable parallel backbonedata relaying and MIMO SDMA operations at the same time is feasiblethrough frequency division by using a portion of the millimeter-wavefrequency spectrum (e.g., ˜60 GHz) for SDMA mobile operations and theremaining spectrum for backbone operation. This obviates the need toprovide costly fiber backbone to ferry the data back and forth.

Even with SDMA, the number of mobile units that the network can supportsimultaneously is still fairly limited, bearing in mind that a N×M MIMO(N is the number of network nodes in LOS of a mobile unit, and M is thenumber of discrete antennas for the mobile unit) can support at mostmin(N, M) simultaneous conversations within the same frequency band. Forexample, if M is 6 to provide true omnidirectional communications, andN>>M, then at most 6 mobile users can use the virtual wires at the sametime. However, because it is theoretically possible to provide up to 100Gbps throughput for a network-mobile link, it would be feasible toprovide additional TDMA-type channel sharing by rapidly delivering alarge data load to one mobile unit, and then redirect the beam todeliver another jumble data packet to another mobile user, and so on,and so forth. Beam hopping is far more power efficient than thetraditional FDMA/CDMA/TDMA approach to multiple access for the simplereason that to achieve the same receiver SNR, a narrow spot beamrequires far less total power because most of the radiated power isdirected at the receiver, so there is no wasted power irradiating atlocations which are far from where the intended recipient is. Thereverse situation where a mobile unit is transmitting to the network isalso true even though the reason for high power efficiency is not asobvious. In the reverse case, the network nodes form a giant coherentphased radiating array with extremely high effective antenna gain, whichallows the mobile unit to use much lower power as the receiving aperturearray greatly amplifies the weak signal from the mobile unit.

In the beam hopping scenario, the individual network nodes must be ableto precisely determine the direction of the mobile unit in relation tothe node itself for beam hopping to work. Inn some embodiments, thedetermination of the direction is accomplished by estimating the arrivalangle of the signal from the mobile unit, which amounts to comparing therelative phases among a plurality of phased array antennas and use therelative phases to back track the mobile signal path. However, since atthis point the node still does not know the direction of the mobilesignal path, hence, by definition, the node antennas are not properlyphased to detect the weak signal from the mobile unit (the assumptionhere is that the mobile unit always initiate the communications througha link connect request), therefore the node might not be able to detectthe mobile unit's connect request. However, this can be solved by makingthe mobile link request a long (1 microsecond) pure tone to the network.Since typical symbol length is around 1 nanosecond, a microsecond toneis 3 orders of magnitude longer, which can be picked up easily by usinga low pass filter (an integrator) which essentially amplifies the signalby a factor of 1000 or 30 dB for a net gain in SNR of 15 dB.

In some embodiments, the out-of-band control plane overlay network isused to monitor the mobile link request (assuming the mobile unit isalso equipped with a low band radio) and relay its GPS locationinformation to the relevant nodes and at the same time informs themobile unit to aim its antennas toward the most visible node. FIG. 10 isa diagram 1000 illustrating an example of an out-ot-band (OOB) toMU-MIMO connection according to some embodiments, where multiple users1004 a and 1004 b are connecting to a repeater network including a node1002 a connected to a gateway and other nodes 1002 b connected to thenode 1002 a.

In a specific implementation, out-of-band (OOB) control channels areused in addition to GPS to locate a user once the user makes a linkrequest. The location information is sent to all nodes of thecommunication network. A channel matrix is computed from location datafor multiple users and eigen-vectors are computed. Location data alonemay not determine phases accurately because a few mm error can changephase. Phase estimation will be discussed later.

For TX, the eigen-coefficients are sent to nodes to “pre-code” data toachieve orthogonal transmission for each user. For RX, the sameeigen-coefficients are used. Starting from node 1, each node performs a“multiply and accumulate” operation taking the data sent from the user,multiplying it with the coefficient and adding to the data sent from theprevious node, and passing that to the next node. This provides theoptimal MIMO decoding for multiple users. This procedure requires thenodes to be ordered, though exact ordering does not matter.

In a specific implementation, a sub-optimal greedy algorithm may beemployed. According to the sub-optimal greedy algorithm, each node usesbeam-forming to steer multiple beams to multiple users. Again, thechannel matrix is computed and sent to each node via the controlchannels. A time delay is introduced to each beam according to thedistance between the node and the user. Each beam is also phase-shiftedto ensure all beams from different nodes will arrive with the exactlysame phase. The greedy algorithm should work about as well as the fullMIMO approach when users are not with a beam width of one another.

In some embodiments, once a node steers its antennas in a rightdirection, proper arrival angle estimation can be performed to furtherrefine the aim. The more accurate directional information can then beconveyed to other visible nodes and to the mobile unit to improve theiraims.

Indoor penetration is yet another major issue with the millimeter-wavefrequency range (e.g., ˜60 GHz). It is possible to install repeaterindoors to alleviate the issue. However, a typical indoor environmentmay contain too many obstacles that impede LOS communication. In orderto address this issue, in some embodiments, the repeater network is usedfor a high-speed data backbone and the overlay lower-frequency meshnetwork is used to provide indoor connection. Since IEEE 802.11ac radiofrequency signals support near gigabit speed in its MIMO mode, this is areasonable solution.

6. Phase Difference

One way to obtain the phase difference between two nodes is to employ apair of direct repeaters for these two nodes. The first node sends acontinuous pure tone of certain duration, and the second node phaselocks its local oscillator (VCO) to the incoming pure tone, and sends anamplified signal back. The first node then compares the phase differencebetween the original signal and the received signal from the secondrepeater. The phase difference between the two nodes is simply the phasedifference observed by the first repeater divided by 2. This wouldrequire the first repeater to both transmit and receive signals with thesame frequency. In this case it would be difficult to maintain a 100 dBisolation between TX and RX.

To address this issue, in an alternative, the second node up-convertsthe received signal by a predetermined amount (e.g., 2 GHz), and sendsthe up-converted pure tone back. Upon receiving the transponded signalfrom the second node, the first node can down-convert the receivedsignal by the predetermined amount (i.e., 2 GHz), and then compute thephase difference between the original signal and the receiveddown-converted signal. The phase difference between the two nodes isthen determined by dividing the computed phase difference by 2. Usingtransponders instead of repeaters can overcome the issue of isolationdiscussed above.

FIG. 12 is a flowchart 1200 of an example of a method of using a controlplane network to provide network dimensional parameters to communicationnetwork nodes. In the example of FIG. 12 , the flowchart 1200 starts atmodule 1202 with receiving a first signal from a first node and a secondsignal from a second node, wherein the second signal is are-transmission of the first signal. In a specific implementation, thefirst node and the second node are positioned along a line andpositioned with other nodes to form a repetitive pattern. Because thepositioning is repetitive, computations during signal processing can besimplified. In a specific implementation, the first signal and thesecond signal are millimeter-wave signals in a frequency between 20 GHzand 120 GHz. In a specific implementation, the first signal issynchronized to a GPS clock signal and the second signal is synchronizedto the first signal and/or to the GPS clock signal. In a specificimplementation, the first signal (and/or second signal) is received overa primary wireless link when the primary wireless link is available andover a secondary wireless link when the primary wireless link isunavailable.

In the example of FIG. 12 , the flowchart 1200 continues to module 1204with receiving network dimensional parameters from one or more controlplane network nodes. Control plane network nodes can be co-located withone or more communication network nodes (e.g., on the same device).Techniques described in this paper can be utilized if relative positionof communication network nodes can be determined with accuracy that isslightly better than that enabled by GPS (e.g., within cm). A controlplane can be used to send network dimensional parameters, which can bereal-time or time-varying.

In a specific implementation, the first node transmits the first signalover a current generation network, the second node transmits the secondsignal over the current generation network, and the one or more controlplane nodes transmit the network dimensional parameters over a previousgeneration network, and wherein the network dimensional parametersinclude real-time or time-varying coefficients. For example, if thecontrol plane is a 4G network, the communication network may be 5G.Better control over physical placement of nodes reduces the amount ofcontrol plane information that is needed for improving network efficacy,so if the communication network is upgraded to 6G in the future, it maynot be necessary to utilizing the older 5G network (the 4G network maystill be considered adequate). The 4G communication network could alsobe used with a 3G network control plane, but the 3G network could beconsidered inadequate for meeting desired benchmarks. The difficultiesincrease as the repetitive pattern of network nodes increases incomplexity (from a line to a square, to a rectangle, to a hexagon, andso forth).

In the example of FIG. 12 , the flowchart 1200 continues to module 1206with correlating the first signal and the second signal as a function ofone or more of the network dimensional parameters. In a specificimplementation, the function uses three network dimensional parameters:respective distances of the first node and the second node in therepetitive placement pattern or values from which the respectivedistances can be determined, signal processing delay at the first nodeand the second node or values from which signal processing delay can bedetermined, and attenuation associated with the first signaltransmission and the second signal transmission or values from whichattenuation can be determined.

In the example of FIG. 12 , the flowchart 1200 continues to decisionpoint 1208 where it is determined whether correlation converges. Ifcorrelation does not converge (1208-N), then the flowchart 1200continues to module 1210 with applying an algorithm for convergence andreturns to module 1206 as described previously. One or more additionalnetwork dimensional parameters may or may not be received and used (see,e.g., module 1204) between a first iteration of module 1206 and a seconditeration of module 1206.

If, after zero or more iterations of module 1210, correlation converges(1208-Y), then the flowchart 1200 ends at module 1212 with transmittinga simplified signal corresponding to correlation of the first signal andthe second signal. In a specific implementation, the third nodeco-phases the first signal and the second signal to generate thesimplified signal. In a specific implementation, the simplified signalis a digital infinite impulse response (IIR) filtered signal implementedin the digital domain after analog-to-digital conversion (ADC).

These and other examples provided in this paper are intended toillustrate but not necessarily to limit the described implementation. Asused herein, the term “implementation” means an implementation thatserves to illustrate by way of example but not limitation. Thetechniques described in the preceding text and figures can be mixed andmatched as circumstances demand to produce alternative implementations.

Furthermore, the repeater network of the present disclosure isdistinguishable over prior art at least as follows. Hypothetically, athird node has a received signal which is a composite of a signal from afirst node, twice delayed and attenuated, and a signal from a secondnode with signal delay and attenuation. Hence the received signal at athird node is greatly distorted. Subsequent nodes following the thirdnode therefore receive progressively more distorted signals. Suchdistortion has to be signal-processed using intra-node echo cancellationand similar techniques to partially compensate for such distortion.However, none of these signal processing techniques work effectively forlater nodes in a long linear signal chain owing to the accumulateddelay/distortion where the combination of larger and larger delayspread, attenuation, and even signal processing distort the receivedsignal so much that no intra-node signal processing technique restoresthe signal effectively.

The disclosure takes advantage of the fact that, given nodes withsufficient co-linearity and approximate equidistant positioning, acollinear network chain is considered a wireless tap-delayed line, andall the tricks which are previously applied to a single board discreteactive RF (radio frequency) filter implementation or to a single chipMMIC (microwave monolithic Integrated Circuit) or RFIC (radio frequencyintegrated circuit) design are utilized to turn the collinear node chaininto a FIR or IIR tap-delayed filter. This solves any delay-spreadinduced signal distortion, without needing any sophisticated adaptiveFIR signal processing at each node. Note that ideally, a completecancellation of delayed signals from multiple nodes is only possiblewith an IIR filter, and such filtering must involve multiple nodes, sothe only alternative is to inter-connect all the nodes within a singlechain with a “LONG” linear waveguide with each node serving as a tap.Doing this however, precludes the need the wireless network in the firstplace.

Other tricks include an IIR inverse sinc filter to minimize theprominent sinc noise generated by the ADC 802. This technique again isimplemented on a single board or at the least on a single device, ratherthan across multiple physically spaced apart nodes. This technique isparticularly useful with ADC implementation.

Control plane and controller design are strictly of the in-band naturewhere the local controller communicates with other service nodes via anetwork interface unit for strictly in-band communication. In-bandherein means that the high bandwidth service data and extremely lowbandwidth control data share the same transport channel with Ethernetchannels. Therefore, the controller is virtualized. However, even avirtualized controller which is software-emulated on another, morepowerful computer, still uses the same local area network thru itsvirtualized network interface, so it is still an “in-band” design.

Returning to FIG. 8 , the disclosure specifically employs an out-of-bandcontrol plane where each node has a non-shared non-virtualizedcontroller on a highly redundant low-frequency (long range) secondarywireless mesh network as an overlay control plane. Such a network has amuch higher degree of inter-connectivity and resilience than the mainmesh network. This permits the control plane to maintain fullconnectivity even when the digital-IIR repeater mesh is severelyimpaired. Since the main repeater network relies on the control plane todetect network faults and arrange rerouting control information toinstruct remaining un-impaired nodes to perform the necessary reroutingoperations to “self-heal”, the control plane is more robust in resistingany disruption of connectivity of the control plane signaling.

A self-organizing mesh network operating in the IEEE802.11ad mmWaveenvironment addresses the issue of enhancing the 802.11ad standard topermit multi-hopping. To enable a multi-hop, each 802.11ad node isconnected optically through fast wired links so that control informationand instructions are sent through an external centralized controller tocoordinate, and schedule the wireless nodes to transfer data packetsbetween a pair of wireless nodes of source node S and destination nodeD, or multi-hopping from a wireless source node as a relay to a remotedestination node.

In an embodiment of the disclosure, a control plane is out-of-band, asin an optical network. A wireless mesh is essentially a residential(last 30 feet) network, based on a multi-hop feature extended to morethan a few hops without suffering major reduction in availablethroughput. In a further embodiment, the disclosed network supportsthousands of hops without any perceptible reduction in overallthroughput owing to an inherently super-low latency (less than 1 nsdepending only on ADC latency) and distortion canceling effect of thedisclosed spatial digital IIR processing.

Some embodiments include an ADC within a transceiver chip based on anADC being a necessary component of any RF baseband signaling. A Balundesign provides impedance matching between a single ended circuit suchas an antenna and a differential circuit such as a coax cable, or a RFfront end, and so on. To down convert the RF signal to baseband linecode/digital signal, an ADC must be used.

In other embodiments of the disclosure, an ADC takes advantage of thehigh performance of the digital IIR filter implementation and its smallsize. An analogue implementation of a sufficiently high order IIRprocessing is unnecessarily bulky and expensive. Therefore, the additionof a low end ADC to the repeater takes advantage of the built-in FEC(forward error correction) coding of the mm-Wave signal to “un-distort”the raw received signal, further enhancing the IIR performance. Note,the preferred embodiment is not to perform any ADC conversion in orderto reduce delay spread, so that large delay spread only occurs for laternodes. In the present disclosure, the digital IIR is sufficient toalmost completely eliminate the delay spread induced distortion, sounwanted delayed signals don't accumulate.

Meticulous design steps are taken to acquire highly accurate carrierphase information since the entire disclosed collinear chain is almostcompletely phase coherent so that each node can act as a node of a verylarge scale linear phased array. In some instances, only a GPS receiveris used or a local timer synched to a GPS receiver is needed. There areno PLL (phase locked loop) and no phase synchronization requirementsbetween TX and RX within the same node. Also, no iterated linear searchwith beacon signaling to fully “phase synchronize” the entire linearchain is required so a GPS time stamp is just that, to time stamp anobserved sensor event for posterity. Such time stamping only needs to beaccurate up to the second, if even that! In the event one wishes to havea much more accurate time stamp, a mechanism to do so is provided asreserved to further development of the present disclosure.

Notwithstanding, time stamp synchronization is distinguishable fromcarrier phase synchronization in the disclosure. The difference issimilar in scope to that between the timing of a strobe light and thephase modulation of a laser beam. A narrow band laser light is highlyphase coherent, whereas a strobe light is emitted from a broad bandlight source as almost all light sources except laser light sourcesdon't even have a well defined phase. Therefore, phase modulating abroad band light source is difficult to achieve, and only amplitudemodulation is readily possible per the strobe light.

The secondary wireless network is an overlay network always working toprovide long range, highly resilient and high connectivity (higher thanthe primary network by many folds) low bandwidth communications for thecontrol plane signaling. The control plane signaling does not go thruthe primary network unless one uses it as a backup for the controlnetwork in the extreme case when the highly robust secondary network isshutdown for exceptional and rare cases.

Embodiments of the disclosure specifically use an out-of-band controlplane, meaning each node has a non-shared non-virtualized controllerwhich uses a highly redundant low-frequency and long range secondarywireless mesh network as an overlay control plane. Such a network has amuch higher degree of inter-connectivity and resilience than the mainmesh network. This permits the control plane to maintain fullconnectivity.

Accordingly, it is not intended that the disclosure be limited, exceptas by the specification and claims set forth herein. While the foregoingis directed to embodiments of the present invention, other and furtherembodiments of the invention may be devised without departing from thebasic scope of the disclosure.

Although the operations of the method(s) herein are shown and describedin a particular order, the order of the operations of each method may bealtered so that certain operations may be performed in an inverse orderor so that certain operations may be performed, at least in part,concurrently with other operations. In another embodiment, instructionsor sub-operations of distinct operations may be implemented in anintermittent and/or alternating manner.

We claim:
 1. A system comprising: a primary plurality of nodes having arepetitive placement pattern, including a first node, a second node anda third node coupled to the first node and to the second node via awireless medium and one or more control plane network nodes coupled tothe third node via the wireless medium; and a secondary wireless meshnetwork of nodes comprising a redundant low-frequency long rangeout-of-band control plane where each node thereof has a non-sharednon-virtualized controller acting as an overlay control plane for theprimary plurality of nodes; wherein the third node transmits a digitalinfinite impulse response (IIR) filtered signal corresponding tocorrelation of the first signal and the second signal to one or more ofthe plurality of nodes, the digital IIR filtered signal implemented in adigital domain after analog-to-digital conversion (ADC) of the first andsecond signals, and wherein the first node transmits the first signalover a current generation network, the second node transmits the secondsignal over the current generation network, and the one or more controlplane network nodes transmit the network dimensional parameters over aprevious generation network, and wherein the network dimensionalparameters include real-time or time-varying coefficients.
 2. The systemof claim 1, wherein the first, second, and third nodes are aligned alonga linear signal path and the network dimensional parameters enable thethird node to determine a precise location of the first node and of thesecond node.
 3. The system of claim 1, wherein the first signal and thesecond signal are millimeter-wave signals in a frequency between 20 GHzand 120 GHz.
 4. The system of claim 1, wherein the third node co-phasesthe first signal and the second signal to generate the digital IIRfiltered signal.
 5. The system of claim 1, wherein the third nodeincludes a reception module configured to perform analog-to-digitalconversion (ADC) and a transmission module configured to generate thedigital IIR filtered signal, and the reception module and thetransmission module are provided on different substrates.
 6. The systemof claim 1, wherein the first node is configured to generate the firstsignal in synchronization with a global positioning system (GPS) clocksignal, and the second node is configured to generate the second signalin synchronization with the GPS clock signal.
 7. The system of claim 1,the first node transmits the first signal over a primary wirelessfrequency range associated with a primary wireless link when the primarywireless link is available, and through a secondary wireless frequencyrange associated with a secondary wireless link when the primarywireless link is unavailable.
 8. The system of claim 1, wherein thefirst node generates the first signal from a downlink signal from agateway node and through a set of intervening nodes.
 9. The system ofclaim 1, wherein the first node generates the first signal from anuplink signal from a wireless mobile unit coupled to the first nodethrough a set of intervening nodes.
 10. The system of claim 1, whereinthe network dimensional parameters include values indicating respectivedistances of the first node and the second node in the repetitiveplacement pattern or values indicating signal processing delay at thefirst node and the second node or values indicating attenuationassociated with the first signal transmission and the second signaltransmission.
 11. A method comprising: operating a primary plurality ofnodes having a repetitive placement pattern, including a first node, asecond node and a third node coupled to the first node and to the secondnode via a wireless medium and one or more control plane network nodescoupled to the third node via the wireless medium; overlaying asecondary wireless mesh network (SWMN) of nodes onto the primaryplurality of nodes, the SWMN comprising a redundant low-frequency longrange out-of-band control plane where each node thereof has a non-sharednon-virtualized controller acting as an overlay control plane for theprimary plurality of nodes; transmitting by the third node, a digitalinfinite impulse response (IIR) filtered signal corresponding tocorrelation of the first signal and the second signal to one or more ofthe plurality of nodes, the digital IIR filtered signal implemented in adigital domain after analog-to-digital conversion (ADC) of the first andsecond signals, wherein the first node transmits the first signal over acurrent generation network, the second node transmits the second signalover the current generation network, and the one or more control planenetwork nodes transmit the network dimensional parameters over aprevious generation network, and wherein the network dimensionalparameters include real-time or time-varying coefficients.
 12. Themethod of claim 11, further comprising sending control information andinstructions through an external centralized controller to coordinateand schedule the wireless nodes to transfer data packets between a pairof wireless source nodes and destination nodes.
 13. The method of claim11, wherein a control plane signaling goes through the SWMN and throughthe primary plurality of nodes as a backup for the overlay control planein an extreme case when the SWMN is shutdown.
 14. The method of claim11, wherein the third node co-phases the first signal and the secondsignal to generate the digital IIR filtered signal and the networkdimensional parameters enable the third node to determine a preciselocation of the first node and of the second node.
 15. The method ofclaim 11, further comprising: synchronizing the first signal to a globalpositioning system (GPS) clock signal; and synchronizing the secondsignal to the GPS clock signal.
 16. The method of claim 11, furthercomprising: receiving the first signal over a primary wireless link whenthe primary wireless link is available; and receiving the first signalover a secondary wireless link when the primary wireless link isunavailable.
 17. The method of claim 11, wherein the network dimensionalparameters include values indicating respective distances of the firstnode and the second node in the repetitive placement pattern or valuesindicating signal processing delay at the first node and the second nodeor values indicating attenuation associated with the first signaltransmission and the second signal transmission.
 18. A systemcomprising: a module configured to operate a primary plurality of nodeshaving a repetitive placement pattern, including a first node, a secondnode and a third node coupled to the first node and to the second nodevia a wireless medium and one or more control plane network nodescoupled to the third node via the wireless medium; a secondary wirelessmesh network (SWMN) module configured to overlay nodes thereof onto theprimary plurality of nodes, the SWMN comprising a redundantlow-frequency long range out-of-band control plane where each nodethereof has a non-shared non-virtualized controller acting as an overlaycontrol plane for the primary plurality of nodes; a module configured totransmit a means for transmitting, by the third node, a digital infiniteimpulse response (IIR) filtered signal corresponding to correlation ofthe first signal and the second signal to one or more of the pluralityof nodes, the digital IIR filtered signal implemented in a digitaldomain after analog-to-digital conversion (ADC) of the first and secondsignals, wherein the first node transmits the first signal over acurrent generation network, the second node transmits the second signalover the current generation network, and the one or more control planenetwork nodes transmit the network dimensional parameters over aprevious generation network, and wherein the network dimensionalparameters include real-time or time-varying coefficients.
 19. Thesystem of claim 18, further comprising an external centralizedcontroller configured to send control information and instructions tocoordinate and schedule the wireless nodes to transfer data packets viamulti-hopping from a wireless source node as a relay to a remotedestination node.
 20. The system of claim 18, wherein the SWMN module isfurther configured with sufficient data transport capacity to relayinformation to mesh network nodes for real-time or time-varyingcoefficients and other network dimensional parameters enabling arelatively precise determination of the location of any mesh networknodes.